This calculator estimates the small-signal -3 dB bandwidth using two common limitations: electrical RC loading and carrier transit time.
- Ctotal = Cj + Cp
- Req = Rs + (RL || Rin)
- fRC = 1 / (2π Req Ctotal)
- ftt ≈ 0.45 / ttr where ttr is the carrier transit time.
To combine two independent limits, the default method uses a quadrature rule:
1 / ftotal2 = 1 / fRC2 + 1 / ftt2
Rise time is estimated by tr ≈ 0.35 / BW. These relationships are approximations that assume a single-pole response near the cutoff.
- Enter the junction capacitance and any expected parasitic capacitance.
- Provide the load resistance and series resistance from wiring or bias networks.
- If an amplifier is used, enable input resistance to model loading accurately.
- Choose a transit-time input method: direct time, or width with carrier velocity.
- Click Calculate Bandwidth to see RC, transit, and total estimates.
- Use Download CSV for records or Download PDF for sharing.
| Case | Cj (pF) | Cp (pF) | Rload (Ω) | Rs (Ω) | ttr (ns) | Estimated BW (MHz) |
|---|---|---|---|---|---|---|
| Fast receiver | 0.5 | 0.2 | 50 | 1 | 1.0 | ~447.725 |
| Higher capacitance | 2.0 | 0.5 | 50 | 2 | 2.0 | ~221.294 |
| Transit limited | 0.3 | 0.2 | 50 | 1 | 5.0 | ~89.991 |
1) Bandwidth as a system requirement
Photodiode bandwidth is the small-signal frequency range where the electrical output still follows optical modulation. For pulse systems, it also relates to edge speed through rise time. This calculator turns capacitance, resistance, and transit assumptions into a practical bandwidth estimate for early design decisions.
2) RC loading usually dominates first
The electrical pole is set by total capacitance and the effective resistance seen by that capacitance: f = 1/(2πRC). As a data point, 50 Ω with 1 pF gives about 3.18 GHz, while 0.5 pF gives about 6.37 GHz. Small parasitics can matter a lot at high speed.
3) Typical parameter ranges you will see
Many small-area photodiodes have junction capacitance in the 0.1 to 2 pF range at useful reverse bias, while larger devices can be several pF. Parasitic capacitance from packages, pads, and PCB routing commonly adds 0.1 to 0.5 pF. Series resistance may be a few ohms, but wiring can add more.
4) Load choices trade bandwidth for signal level
A 50 Ω termination improves bandwidth but reduces voltage per photocurrent compared with high impedance front ends. If an amplifier input resistance is comparable to your load, it acts in parallel and changes the RC pole. Modeling RL || Rin helps you avoid overestimating speed.
5) Transit time limits the intrinsic device speed
Even if RC is excellent, carriers still need time to cross the depletion region. A common approximation is f ≈ 0.45/t. For example, 100 ps corresponds to about 4.5 GHz, while 2 ns corresponds to about 225 MHz. If your transit limit is low, optimizing RC alone will not help.
6) Estimating transit time from geometry
If transit time is unknown, estimate it as t = W/v using depletion width and carrier velocity. A 3 µm depletion width with 1×105 m/s gives about 30 ps, supporting multi-GHz operation if the RC pole is also high.
7) Combining limits and interpreting rise time
Real responses often include both RC and transit mechanisms. The quadrature combination used here treats them as independent limits and produces an overall bandwidth below each individual limit. Rise time is reported using tr ≈ 0.35/BW, giving a time-domain view that is useful for pulses and timing budgets.
8) Practical validation and layout notes
This model is intentionally first-order. At very high speed, packaging inductance, frequency-dependent capacitance, bias networks, and amplifier dynamics can dominate. Keep interconnects short, control impedance, and verify with measurement or simulation. Use the CSV/PDF exports to compare design variants consistently.
1) What does the overall bandwidth represent?
It is a combined -3 dB bandwidth estimate based on an RC limit and a carrier transit-time limit. You can combine them using quadrature (default) or by taking the minimum as a quick dominant-limit check.
2) When should I use the minimum method?
Use minimum when you only need to know which mechanism dominates or you want a conservative screening value. Quadrature is better when both effects are comparable and you want a smoother combined estimate.
3) How do I choose parasitic capacitance?
Include pad capacitance, package capacitance, connector capacitance, and PCB trace capacitance near the diode. If you do not know it, start with 0.2 to 0.5 pF and refine after layout or from vendor package models.
4) Should I include amplifier input resistance?
Yes if the amplifier presents a finite resistive load at the photodiode node. If Rin is large compared with RL, the effect is small. If it is comparable, it can noticeably shift the RC pole.
5) Is the 0.35/BW rise-time rule always valid?
It is a common single-pole approximation for 10 to 90% rise time. If your response has peaking, multiple poles, or equalization, the mapping changes. Treat rise time here as an engineering estimate, not a guaranteed measurement.
6) Can this be used for transimpedance amplifiers?
It can provide a first-order picture if you translate the front-end loading into an effective resistance and capacitance. However, a TIA bandwidth also depends on feedback components, amplifier gain-bandwidth, and stability, which are not explicitly modeled.
7) What important effects are not modeled?
The calculator does not include inductance, transmission-line behavior, frequency-dependent diode capacitance, diffusion tails, or noise. For multi-GHz designs, packaging and PCB parasitics can dominate. Use this tool for sizing, then validate with RF measurement or circuit simulation.